LMS-based adaptive pre-distortion for enhanced power amplifier efficiency

ABSTRACT

Systems and methods are disclosed to adaptively pre-distort a signal prior to being used by a non-linear circuit, such as a higher power amplifier (at transmit) or a low noise amplifier (at receive). The signal is compared with a feedback signal from the non-linear circuit and a metric is calculated and minimized. The input signal is adaptively changed, such as by varying tap coefficients, until the metric is sufficiently minimized, resulting in a pre-distorted signal that is substantially linear upon passing through the non-linear circuit.

TECHNICAL FIELD

This invention relates generally to communication systems and, moreparticularly, to pre-distortion circuitry for use with power amplifiers.

BACKGROUND

Many types of communication systems use a power amplifier as part of thefront end of a transmitter for transmitting RF signals. In some systems,a low noise amplifier (LNA) at the front end of a receiver may be usedfor receiving RF signals. However, a power amplifier has non-lineardistorting characteristics that can cause distortion of the informationsignal being amplified. The non-linear distorting characteristics of thepower amplifier can impact the instantaneous amplitude and phase of thesignal significantly. Furthermore, non-linearity within the RFcomponents can introduce distortion in the signal and reduced SNR orpoorer performance at the receiver. In wireless systems, non-linearityin the power amplifier at transmission can introduce intermodulationdistortion which can result in spectral emissions in the adjacentchannels. Typically, tight specifications in terms of spectral masks areprovided by Standards Committees and/or Government bodies, which put amaximum limit to such spurious, out-of-band emissions for complianttransceivers. Therefore, it is desirable to provide a linear signal outof the power amplifier.

One type of method to linearize the power amplifier output signal is to“pre-correct” the signal being input to the amplifier, also known aspre-distortion. There are many known techniques are used to pre-correctan information signal in order to linearize the output of the amplifier.One of these techniques involves amplitude correction which produces alinear piece-wise pre-correction function which is correlated to thenon-linear characteristics of the amplifier. The result is a piece-wisecorrection curve which approximates the ideal correction. The correctionis then added to the information signal.

Pre-distortion has been almost exclusively in the baseband domain. Thetypical approach has been to apply a pre-distortion function at digitalbaseband. A relatively recent but commonly used approach has been tostore the pre-distortion function as a look-up table which stores thegain and phase values as a function of the input signal envelope. Theinput signal is compared with the feedback signal from the poweramplifier and certain metrics, such as ratio of in-band to out-of-bandemission power and/or their correlation are used to adaptively map thelook-up table with respect to the envelope of the input signal. Thus,the data path is simply a complex multiplication at digital baseband;the adapt-path (or feedback path) consists of a look-up table indexed bythe input signal envelope and the adaptation metric is the ratio ofin-band to out-of-band emission or the correlation between the feedbackand the input signals. These operations are carried out at digitalbaseband.

However, increasing the linearity of the power amplifiers can reduce thepower efficiency of the amplifiers, make them more-voluminous, requiremore cooling equipment, and substantially increase the cost andform-factor of the transceivers. To improve the linearity of thetransmission without resorting to a higher end and more expensive poweramplifier, one method that has been carried out in the prior art is toperform pre-distortion on the transmitted signal before the signal isinput into the power amplifier such that the pre-distortion equalizes insome sense the non-linear post-distortion of the amplifier. Thepre-distortion may be carried out within an integrated circuit; however,a significant challenge has always been to be able to adaptively obtainthe pre-distortion transfer function so as to be optimal in some metric.It is also desirable to have a tracking mechanism with thepre-distortion so that different temperature and aging effects are alsocompensated for.

Accordingly, it would be desirable to have systems and methods forperforming pre-distortion in power amplifiers that overcome thedisadvantages of the prior art as discussed.

SUMMARY

According to one aspect of the present invention, adaptivepre-distortion is performed using a least-mean square (LMS) method toadaptively obtain an optimal pre-distortion transfer function in themean square error sense. In one embodiment, a LMS-based adaptivepre-distortion circuit coupled between an input signal to be transmittedand a non-linear power amplifier compares the input signal to a feedbacksignal from the power amplifier. The circuit adaptively reduces theleast-mean square error between the two signals using both linear andnon-linear filters and various feedback signals within the circuit. Theresult is an approximately linear output from the power amplifier. Thistechnique can be used with single and multi-carrier transceivers, aswell as with either a power amplifier (at the transmitter) or low noiseamplifier (LNA) (at the receiver). The adaptive pre-distortion can beperformed in the baseband domain or purely in the RF domain, usinganalog continuous-time or discrete-time signal processing.

According to one embodiment, an LMS-based adaptive pre-distortion (LAPD)circuit receives an input signal to be transmitted, such as from abaseband processor and RF circuitry, and a feedback signal from a poweramplifier. The LAPD circuit adaptively minimizes a metric, such as anerror signal, formed by the difference between the two signals. The LAPDcircuit receives the input signal through an adaptive AGC circuit. TheLAPD circuit further includes a feedforward filter, an adaptivecoefficient generator, a slicer, timing control circuit, and sliceroutput time-align circuit. An error timing align circuit within theadaptive coefficient generator may be used to time align the inputsignal from the adaptive AGC circuit with the error signal e(t). Theerror timing align circuit may receive as an input signal a feedback tapcoefficient vector c or the error signal e(t). The timing controlcircuit time aligns the input signal and the feedback signal from thepower amplifier to compute an error signal, which is the time differencebetween the two signals. This error signal is iteratively reduced untila desired signal is obtained. The output signal is then input to thenon-linear power amplifier, which amplifies the signal for transmission,resulting in a linearized output signal.

The scope of the invention is defined by the claims, which areincorporated into this section by reference. A more completeunderstanding of embodiments of the present invention will be affordedto those skilled in the art, as well as a realization of additionaladvantages thereof, by a consideration of the following detaileddescription of one or more embodiments. Reference will be made to theappended sheets of drawings that will first be described briefly.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 shows a block diagram of a portion of a single-carriertransceiver with adaptive pre-distortion according to one aspect of thepresent invention;

FIG. 2 shows a block diagram of a pre-distortion circuit for use in thetransceiver of FIG. 2 according to one embodiment;

FIG. 3 shows a more detailed block diagram of the pre-distortion circuitof FIG. 2 according to one embodiment;

FIG. 3A shows one embodiment of the pre-distortion circuit of FIG. 3with a more detailed feedforward filter and adaptive coefficientgenerator;

FIG. 4 shows one embodiment of a feedforward filter in FIG. 3;

FIG. 5 shows one embodiment of an adaptive coefficient generator of FIG.3;

FIG. 5A shows one embodiment of a controllable delay in FIG. 5;

FIG. 5B shows one embodiment of an interpolation control circuit of FIG.5A;

FIG. 5C shows one embodiment of an interpolation delay circuit of FIG.5A;

FIG. 6 shows one embodiment of a low pass filter block suitable for usein the system of FIG. 5;

FIG. 7 shows one embodiment of a slicer input time align circuit for usein the system of FIG. 3;

FIG. 7A shows one embodiment of an interpolating mixer circuit of FIG.7;

FIG. 7B shows one embodiment of a phase detector circuit of FIG. 7;

FIG. 8 shows one embodiment of a slicer output time align circuit foruse in the system of FIG. 3;

FIG. 8A shows one embodiment of a coefficient generator circuit of FIG.8;

FIG. 8B shows one embodiment of a timing interpolation parameter signalgenerator circuit of FIG. 8;

FIG. 9 shows a block diagram of a multi-carrier transceiver withadaptive pre-distortion according to one embodiment; and

FIG. 10 show a block diagram of a multi-carrier pre-distortion circuitof FIG. 9 according to one embodiment.

Embodiments of the present invention and their advantages are bestunderstood by referring to the detailed description that follows. Itshould be appreciated that like reference numerals are used to identifylike elements illustrated in one or more of the figures.

DETAILED DESCRIPTION

FIG. 1 is a block diagram of a portion of an RF communication system 100for transmitting RF signals. System 100 includes a baseband processor102 as part of a single carrier transceiver that generates signals inbaseband to be transmitted. These signals are input to one or more RFcircuits 104 that translate the signals to the RF domain. A least-meansquare (LMS) based adaptive pre-distortion (LAPD) circuit 106 processesthe desired signal (x1) to be transmitted from RF circuits 104 andfeedback signals (x2) from a single carrier power amplifier 108 togenerate a pre-distorted signal to power amplifier 108. LAPD circuit 106adaptively pre-distorts the input signal such that after travelingthrough power amplifier circuit 108, the signal is linear. Note thatwith conventional systems, LAPD circuit 106 is replace by a look-uptable that compares the two input signals and assigns specific values tothe signal for pre-distortion.

System 100 further includes a power coupler 110 that receives the outputsignal from power amplifier 108 and splits a portion of the input RFsignal into the feedback path and passes the rest of the signal throughto the antenna for transmission. The output signal from power coupler110 is received by an amplitude control or controllable gain circuit112, which then scales the signal so that the signal input into LAPDcircuit 106 is within an admissible range for operation of the LAPDcircuit. Controllable gain circuit 112 feeds the signal back to LAPDcircuit 106 for comparing with the desired signal to be transmitted.Power coupler 110, after receiving the signal from power amplifiercircuit 108, sends the signal to an antenna 114 for transmission. Afteradaptive pre-distortion by LAPD circuit 106, the signal from poweramplifier circuit 108 is essentially linear.

FIG. 2 is a block diagram of LAPD circuit 106 according to oneembodiment of the invention. LAPD circuit 106 includes an adder 200 thatcalculates the difference between the input signal (x1) from RF circuits104 and the feedback signal (x2) from power amplifier circuit 108.Signal x2 will sometimes be referred to as the feedback signal frompower amplifier 108. However, it is recognized that feedback signal x2may actually pass from power amplifier 108 through power coupler 110 andamplitude control circuit 112. Thus, a feedback signal from the poweramplifier or a low noise amplifier (or other non-linear element) can beany signal that passes through one or more circuit elements beforeentering LAPD circuit 106 and does not require a direct signal from thepower amplifier to the LAPD circuit. The output signal e(t) of adder 200(i.e., the difference or error signal) is input to an LAPD adaptpathcircuit 202, which provides adaptive tap coefficients or vector c fromthe error signal e(t) and the input signal x1. Note that as used herein,adaptpath indicates circuitry that adaptively changes a signal. The tapcoefficients c, along with input signal x1, are then input to an LAPDdatapath circuit 204, which generates an pre-distorted signal z(t). LAPDdatapath-circuit 204 pre-distorts the signal based on a set ofconfiguration parameters (or tap weights) which are controlled by LAPDadaptpath circuit 202 based on minimizing a certain metric, such as meansquared amplitude of the input signal into the LAPD and the amplitudecontrol circuit output. The signal is then filtered through a bandpassfilter 206 for transmission to the power amplifier. In one embodiment,bandpass filter 206 is part of LAPD circuit 106, while in anotherembodiment, bandpass filter 206 is external to LAPD circuit 106.

FIG. 3A is a more detailed block diagram of LAPD circuit 106 of FIGS. 1and 2 according to one embodiment. LAPD circuit 106 includes an adaptiveautomatic gain control (AGC) circuit 300 within LAPD adaptpath circuit202, which provides the necessary gain for small signals. Thus, after asignal is output from AGC circuit 300, the signal is within a suitablerange that allows the signal to be properly used throughout the rest ofthe circuit. Adaptive AGCs are known and any such suitable one can beused with the present invention to adaptively provide the proper gain.LAPD circuit 106 also includes a feedforward filter 302, an adaptivecoefficient generator 304, an output signal slicer 306, a timing controlor time-align circuit 308, and a slicer output time-align circuit 310.Within LAPD circuit 106 is LAPD adaptpath circuit 202, which includesadaptive coefficient generator 304, an adder 318, and an integrator,such as a low pass filter block 316, and LAPD datapath circuit 204,which includes adaptive AGC circuit 300, feedforward filter 302, timingcontrol circuit 308, output signal slicer 306, slicer output time-aligncircuit 310, an adder 312, and a multiplier 314.

Input signal x1 is received by adaptive AGC circuit 300. Feedforwardfilter 302 receives the output signal x1 from adaptive AGC circuit 300and tap coefficients from adaptive coefficient generator 304 andgenerates a pre-distorted signal, which is input to adder 312. The otherinput to adder 312 is the product 314 of the output of low pass filterblock 316, and slicer output time-align circuit 310.

The feedback signal (from product 314) into adder 312 provides aniterative correction to an error signal e(t) for use by adaptivecoefficient generator 304 to generate adaptive tap coefficients. Theerror signal, processing through adder 318, is the difference betweenthe output of timing control circuit 308 and input signal x2 (thefeedback signal from the power amplifier). As time passes, the errorsignal converges until a sufficiently small error signal is obtainedthrough adaptively changing the tap coefficients.

FIG. 3B is a more detailed block diagram of LAPD circuit 106 accordingto one embodiment, in which adaptive coefficient generator 304 andfeedforward filter 302 are shown in greater detail. Details will bediscussed further below.

FIG. 4 shows one embodiment of a feedforward filter suitable for use asfeedforward filter 302 of FIG. 3. The feedforward filter includes aseries of signal delay elements 402-1 to 402-N. Each delay element 402delays the incoming signal by a fixed amount τ, e.g., x1′(t−τ),x1′(t−2τ), . . . x1′(t−Nτ). The delay τis typically selected to be lessthan a symbol period, and in one embodiment, is based on the symbolperiod τ, and the number of feedforward taps N as follows:$\tau = \frac{T_{s}}{N - 1}$The input data signal x1′(t) and each successive delayed signal fromdelay elements 402-1 to 402-N are multiplied by multipliers 404-1 to404-N with its respective adaptive coefficient signals from adaptivecoefficient generator 304. The product signals are then summed by anadder circuit 406 to form the pre-distorted signal.

FIG. 5 shows one embodiment of an adaptive coefficient generator for useas adaptive coefficient generator 304 of FIG. 3. The generator includesa controllable delay 500, which receives input data signal x1′(t) andthe error signal e(t), and introduces a fixed delay into the signal. Onetype of controllable delay suitable for the present invention is anerror timing control (ETC) and precursor/postcursor control (PPC)circuit, such as described in commonly-owned U.S. patent applicationSer. No. 10/290,993, filed Nov. 8, 2002, entitled “Adaptive SignalEqualizer with Adaptive Error Timing and Precursor/PostcursorConfiguration Control”, which is incorporated herein by reference in itsentirety. The controllable delay in the present invention is used toalign the error signal e(t) with the input signal x1′(t). In oneembodiment, the ETC/PPC circuit of the above referenced application isused to adaptively set the delay by using the error signal e(t) as acontrol input signal.

FIG. 5A shows one embodiment of controllable delay 500. Controllabledelay 500 includes an interpolation control stage 552 and aninterpolation delay stage 554. Interpolation control 552 processes theerror signal coefficients to produce a set of delay interpolationcontrol signals for interpolation stage 554. In response to these delayinterpolation control signals, interpolation delay stage 554 processesits input signal x1′(t) to produce the corresponding delayed signal forprocessing by delay elements 502.

FIG. 5B shows one embodiment of interpolation controller 552.Interpolation controller 552 includes a set of signal multipliers 556-1to 556-N, a signal combining circuit 558, a signal integration circuit(e.g., a low pass filter) 560, and a signal complement circuit 562,interconnected substantially as shown. Each of the error signalcoefficient signals e₁ to e_(N) is multiplied in a respective multiplier556-1 to 556-N with a corresponding weighted or scaled signal K₁ toK_(N). In one embodiment, K₁ to K_(N/2) are equal to +1, whileK_((N/2)+1) to K_(N) are equal to −1. The resulting product signals aresummed in signal combiner 558. The sum signal is integrated by signalintegrator 560 to produce the primary delay interpolation control signalrepresenting the timing control ratio parameter r. This delayinterpolation control signal is also complemented by signal complementcircuit 562 to provide the complement delay interpolation controlsignal. Signal complement circuit 562 processes the delay interpolationcontrol signal by subtracting it from a normalized value (e.g., unity)to produce the complement signal. The uncomplemented and complementedsignals are then processed by interpolation delay 554.

FIG. 5C shows one embodiment of interpolation delay 554. Interpolationdelay 554 includes three signal delay elements 572-1, 572-2, and 572-3.The incoming signal, i.e., the input signal x1′(t), and thecorresponding successively time-delayed versions are multiplied insignal multipliers 574-1, 574-2, 574-3, and 574-4 with correspondinginterpolation control signals. The resulting product signals are summedin a signal combiner 576 to produce the delayed version of the incomingsignal.

Referring to FIG. 5, the aligned output signal of controllable delay 500is processed through a series of delay elements 502-1 through 502-N.Delay elements 502 introduce a delay of τ′ to its input signal. In oneembodiment, τ′ is greater than the delay τ of delay elements 402 of FIG.4, although it may also be suitable for τ′ to be approximately equal toτ in other embodiments. Having τ′>τ results in a generally more robustsystem. The delayed signals (by multiples of τ′) are input to respectivelow pass filter blocks 504-1 to 504-N, along with the error signal e(t)from adder 318 of FIG. 3. Low pass filter blocks 504 multiply eachdelayed signal with a corresponding error signal and integrate theresult to generate individual tap coefficient signals for use bymultipliers 404-1 to 404-N of FIG. 4. FIG. 6 shows one embodiment of lowpass filter block 504, which includes a multiplier circuit 600 and anintegrator circuit, such as a low pass filter 602.

In one embodiment, low pass filters 602 are analog (or continuous-time)first-order low-pass filters having a transfer function${H(s)} = \frac{G}{1 + {s \cdot T_{l}}}$where G is the gain of the low pass filter and T₁ is the leakage timeconstant. The filter parameters gain G and time constant T₁ can bechosen to meet system and component requirements. For example, the timeconstant T₁ is selected as a non-zero positive number for more robustperformance with a fractionally-spaced feedforward filter at thefront-end. The gain G should be large enough so that the mismatch of thetap coefficient with the least-mean square value at convergence issufficiently small. Further, T₁>0 and G need to be moderate enough tominimize the effects of “tap coefficient drift”. Another factor inselecting gain G and time constant T₁ is to achieve a convergence timethat is sufficiently small for the system.

Selecting values for G and T₁ depend on system requirements and includefactors such as the maximum steady state mismatch of signals, timevariability of the channel, amount of noise within the channel, andparasitic effects of the circuit. In one embodiment, the gain G may beset to 10 to 50 which will result in small mismatch and possibly stableoperation. The time constant T₁ is typically set to about 10,000 to100,000 symbol times, in one embodiment. For very fast varying channels,T₁ is much smaller. T₁ is larger for channels having less temporalvariations and more noise. Higher order low pass filters are alsosuitable for use with the present invention.

Referring back to FIG. 3, similarly, low pass filter block 316 receiveserror signal e(t) from adder 318 and the output of slicer outputtime-align circuit 310. These two signals are then multiplied andintegrated by low pass filter block 316.

FIG. 7 shows one embodiment of a timing control circuit for use astiming control circuit 308 of FIG. 3. The circuit includes acontrollable delay, which receives the input data signal x1(t) andintroduces a delay Δ into the signal. One type of controllable delaysuitable for the present invention is an adaptive least meansquare-based timing interpolation (ALTI) circuit, such as described incommonly-owned U.S. patent application Ser. No. 10/321,893, filed Dec.17, 2002, entitled “Adaptive Signal Latency Control for CommunicationsSystems Signals”, which is incorporated herein by reference in itsentirety.

The controllable delay in the present invention is used to time alignthe input signal x1 with the feedback signal x2 from the poweramplifier. In one embodiment, the ALTI circuit of the above referencedapplication is used to adaptively set the delay to align the signals byusing the feedback signal x2(t) from the power amplifier. The ALTI blockmay be used to time-align the input signal to compute the distortionerror signal which is the difference between the power amplifier outputsignal and the input signal, i.e., the time-align circuit 308 delays itsoutput signal to compensate for delays introduced by processing of theinput signal by the circuit.

FIG. 7 shows one embodiment of timing control circuit 308, whichincludes an interpolating mixer 702, a phase detector 704, and a signalintegrator 706, interconnected substantially as shown. The feedbacksignal x2(t) is compared in signal phase by phase detector 704 with thedelayed signal from interpolating mixer 702. The resulting detectionsignal is integrated by signal integrator 706 (e.g., a low pass filter)to produce an interpolation control signal r(t) for interpolating mixer702.

FIG. 7A shows one embodiment of interpolating mixer 702, which isimplemented as a tapped delay line with correlated tap coefficients. Theinput signal x1(t) is delayed by a signal delay element 712 which is afractional delay element introducing a delay which is less than one datasymbol period in duration. The resulting fractionally delayed signal andthe original input signal x1(t) are mixed (e.g., multiplied) inrespective signal mixers 714-1, 714-2 with respective interpolationcontrol signals representing timing interpolation parameters. The firsttiming interpolation parameter signal is the feedback signal from signalintegrator 706 (FIG. 7). This signal is also complemented by a signalcomplement circuit 718 in which the input signal is subtracted from anormalized value (e.g., unity) to produce the second timinginterpolation parameter signal. The resultant mixed signals are combined(e.g., summed) in a signal combining circuit 716 to produce the delayedsignal x1(t−Δ).

FIG. 7B shows one embodiment of phase detector 704 of FIG. 7, which canbe implemented using a fractional delay element 722, a signal combiningcircuit 724, and a signal mixer 726, interconnected substantially asshown. The delayed signal x1(t−Δ) is further delayed by fractional delayelement 722, after which it is combined with delayed signal x1(t−Δ) insignal combiner 724 such that the further delayed signal is subtractedfrom the input delayed signal x1(t−Δ). The resulting combined signal ismixed (e.g., multiplied) in signal mixer 726 with the feedback signalx2(t) (and a gain constant G_(r), if desired) to produce the phasedetection signal for signal integrator 706 (FIG. 7A).

FIG. 8 shows one embodiment of a slicer output time-align circuit foruse as slicer output time-align circuit 310 of FIG. 3. The circuitincludes a controllable delay, which receives the data signal y(t) fromslicer 306 of FIG. 3 and introduces a delay into the signal. One type ofcircuit suitable for the present invention is a fat tap adaptation (FTA)circuit, such as described in commonly-owned U.S. patent applicationSer. No. 10/322,024, filed Dec. 17, 2002, entitled “Adaptive CoefficientSignal Generator for Adaptive Signal Equalizers with Fractionally-SpacedFeedback”, which is incorporated herein by reference in its entirety.

The controllable delay in the present invention is used to time alignthe decision feedback signal, which is the signal from slicer 306, to bea symbol period delay with respect to the slicer input signal from whichit is cancelled. A method of obtaining this delay in an adaptive manneris by using the FTA block, as described in the above-referencedapplication, which uses the tap coefficients or error signal e(t) as acontrol input signal.

FIG. 8 shows one embodiment of slicer output time-align circuit 310,which includes multiplier circuits 802-1, 802-2 and control signalgenerator circuitry implemented as a coefficient signal generator 804and a timing interpolation parameter signal generator 806, allinterconnected as substantially shown. The adjacent time-delayedfeedback signals, the output signal y(t) of slicer 306 (FIG. 3) and thesignal y(t−δ) delayed by a symbol period δ, are multiplied in theirrespective multiplier circuits 802-1, 802-2 with the error signal e(t).The resulting product signals e(t)y(t) and e(t)y(t−δ) are processed bysignal generator circuits 804, 806. Coefficient signal generator circuit804 provides an adaptation control signal d(t) to timing interpolationparameter signal generator circuit 806, which in return, provides twoother adaptation control signals r(t) and 1−r(t) back to coefficientsignal generator 804. As a result of processing these input signalse(t)y(t), e(t)y(t−δ), r(t), and 1−r(t), coefficient signal generator 804provides the adaptive coefficient signals d(t)r(t) and d(t)[1−r(t)],where signal element d(t) is the weighting factor and signal elementr(t) is the factor indicative of the degree of correlation between theadjacent time-delayed feedback signals y(t) and y(t−δ).

FIG. 8A shows one embodiment of coefficient signal generator 804.Initial product signals y(t) and y(t−δ) are further multiplied inmultiplier circuits 812-1, 812-2 with the adaptation control signalsr(t) and 1−r(t), respectively, from timing interpolation parametersignal generator 806. An additional constant signal μ_(c) can also bemultiplied as part of the product operations or implemented as aconstant scaling factor within the multiplier circuits 812-1, 812-2.

The resulting product signals are summed in a signal summing circuit814. The resulting sum signal is integrated in an integration circuit816 (e.g., a low pass filter) to produce the first adaptation controlsignal d(t). This adaptation control signal d(t), in addition to beingprovided to timing interpolation parameter signal generator 806, ismultiplied within multiplier circuits 818-1, 818-2 with the otheradaptation control signals r(t) and 1−r(t), respectively, provided bytiming interpolation parameter signal generator 806. The product signalsresulting from these multiplication operations are the adaptivecoefficient signals d(t)r(t) and d(t)[1−r(t)].

FIG. 8B shows one embodiment of timing interpolation parameter signalgenerator 806. The initial product signals e(t)y(t) and e(t)y(t−δ) aredifferentially summed in a signal summing circuit 822, where the secondproduct signal e(t)y(t−δ) is subtracted from the first product signale(t)y(t). The resulting difference signal is multiplied in a multipliercircuit 824 with the adaptation control signal d(t) provided bycoefficient signal generator 804. As with the multiplier circuits 812-1,812-2 of coefficient signal generator 604, an additional constant signalμ_(r) can also be used in this multiplication operation or implementedas a constant scaling factor within multiplier circuit 824 operation.

The resulting product signal is integrated by a signal integrationcircuit 826 (e.g., a low pass filter) to produce one of the adaptationcontrol signals r(t) used by coefficient signal generator 804. Thisadaptation control signal r(t) is further processed by a signalcomplement circuit 828, in which the input signal r(t) is subtractedfrom a reference signal having a normalized value, with the resultingdifference signal 1−r(t) serving as the other adaptation control signalused by coefficient signal generator 804. For example, if the value ofthe incoming signal r(t) were considered to have a normalized signalvalue range bounded by the values of zero and one, signal complementcircuit 828 subtracts the incoming signal r(t) from the value of one toproduce the output signal 1−r(t).

The above described embodiment utilizes an adaptive pre-distortioncircuit with feedback circuitry and both linear and non-linear elements.By selecting the appropriate parameters for the low pass filters, theconvergence of the least mean square value can be controlled to preventcoefficient drift of the adaptive tap coefficients. The continuous timeiterative process results in the ability to adaptively change the tapcoefficients for error minimization and provide a linear signal out of anon-linear power amplifier.

While the above description is for a single-carrier transceiver, thepresent invention can also be used with multi-carrier transceivers. FIG.9 shows a block diagram of a transceiver 900, in which multi-carrieradaptive pre-distortion is performed by a multi-carrier LAPD circuit910. In this example, there are two carriers. Transceiver 900 includestwo baseband processors 102-1 and 102-2, each followed by two RFcircuits 104-1 and 104-2, LAPD circuit 910 that receives the outputs x11and x12 from RF circuits 104-1 and 104-2, respectively, and a feedbacksignal x2 from a multi-carrier power amplifier 912 via power coupler110. Amplitude control circuit 112 (not shown) may be coupled betweenpower coupler 110 and LAPD circuit 910 in some embodiments. A linearizedsignal out of power amplifier 912 can be transmitted through antenna114.

FIG. 10 is a block diagram of multi-carrier LAPD circuit 910 of FIG. 9,according to one embodiment. LAPD circuit 910 includes two “branches”,one for each signal at a different carrier. The input signal x11 fromthe first carrier is utilized by a first branch, which includes LAPDadaptpath circuit 202-1 and LAPD datapath circuit 204-1, such asdescribed above with respect to a single carrier LAPD circuit. The inputsignal x12 from the second carrier is utilized by a second branch, whichincludes LAPD adaptpath circuit 202-2 and LAPD datapath circuit 204-2,again as described above with respect to FIGS. 3A and 3B according toone embodiment. The error signal e(t) is input to LAPD adaptpathcircuits 202-1 and 202-2 for adaptive processing, as described above.However, a difference with the single-carrier LAPD circuit is that theerror signal for multi-carrier LAPD 160 is the difference between poweramplifier feedback signal x2 and all the different input signals (x11and x12 in this example), instead of just a single carrier input signalx1. This difference or error signal is determined by an arithmeticcircuit 1002, such as a multi-input adder.

After adaptively generating a pre-distortion signal for each carrier,the output signals from LAPD datapath 204-1 and from LAPD datapath 204-2are passed through bandpass filters 206-1 and 206-2, respectively. Aswith the single-carrier embodiment, bandpass filters 206 may be internalto or external of multi-carrier LAPD circuit 910. The output signalsfrom bandpass filters 206-1 and 206-2 are then summed by an adder 1004to generate a pre-distorted signal z for power amplifier 912. Due toadaptively pre-distorting input signals according to the presentinvention, the output of a non-linear power amplifier can be made linearwith a wider range of factors, as well as at lower cost and size.

In some applications and systems, the power amplifier and/or other RFcomponents may introduce inter-symbol interference (ISI) in addition tonon-linearity. To compensate for both ISI and non-linearity effectsjointly, the present invention can be modified, in part, by using alarger number of taps N′ such that N′*τ>T_(s).

The above-described embodiments of the present invention are merelymeant to be illustrative and not limiting. It will thus be obvious tothose skilled in the art that various changes and modifications may bemade without departing from this invention in its broader aspects. Forexample, although the multi-carrier mode is shown with two carriers, theinvention can also be used with a multi-carrier transceiver having morethan two carriers by modifying the above descriptions accordingly.Further, the above description has focused on continuous-time, RF domainsignal processing; however, the present invention can be extended ormodified for use with discrete-time signal processing architectures aswell. Therefore, the appended claims encompass all such changes andmodifications as fall within the true spirit and scope of thisinvention.

1. A pre-distortion circuit, comprising: an adder coupled to receive afeedback signal from a non-linear power amplifier and a first signal,wherein the adder generates a difference signal between the feedbacksignal and the first signal, wherein the first signal is a signal to betransmitted that is time-aligned with the feedback signal; an adaptivecoefficient generator configured to receive the difference signal and asecond signal, wherein the second signal is a version of the signal tobe transmitted, and wherein the adaptive coefficient generator generatesa set of tap coefficients; and a timing and filter circuit configured toreceive the feedback signal and the signal to be transmitted andgenerate the first signal and receive the tap coefficients and generatea pre-distorted output signal based on a mean square error.
 2. Thecircuit of claim 1, further comprising a bandpass filter coupled toreceive the output signal.
 3. The circuit of claim 1, wherein the timingand filter circuit comprises: a feedforward filter adapted to receivethe adaptive tap coefficients and the second signal to provide a filteroutput signal; a slicer adapted to receive a slicer input signal andprovide a slicer output signal; a slicer timing alignment block adaptedto receive the slicer output signal and provide a second output signal;and a low pass filter adapted to receive the second output signal andthe error signal and provide a third output signal, wherein the thirdoutput signal is multiplied with the second output signal to provide afeedback signal which is added to the filter output signal to generatethe slicer input signal.
 4. The circuit of claim 3, wherein thefeedforward filter comprises a series of N delay elements, wherein eachdelay element introduces a delay ${\tau = \frac{T_{s}}{N - 1}},$ whereT_(s) is the symbol period.
 5. The circuit of claim 3, wherein thefeedforward filter comprises a series of N delay elements, with N beinggreater than τ/T_(s) where τ is the delay introduced by each delayelement and T_(s) is the symbol period.
 6. The circuit of claim 1,wherein the pre-distortion circuit provides a continuous time adaptationfor the power amplifier.
 7. The circuit of claim 1, wherein the adaptivecoefficient generator time-aligns the second signal.
 8. The circuit ofclaim 3, wherein the slicer timing alignment block time-aligns theslicer input signal with the slicer output signal.
 9. The circuit ofclaim 1, further comprising an adaptive automatic gain control circuitconfigured to receive the signal to be transmitted and provides thesecond signal.
 10. The circuit of claim 1, wherein the power amplifieris a single-carrier power amplifier.
 11. The circuit of claim 1, whereinthe power amplifier is a multi-carrier power amplifier.
 12. The circuitof claim 1, further comprising a timing control circuit configured toreceive the signal to be transmitted and the feedback signal and outputsthe first signal.
 13. An RF transceiver, comprising: an adaptivepre-distortion circuit configured to receive an RF signal and a feedbacksignal and adaptively generate a pre-distortion signal; and a non-linearcircuit configured to receive the pre-distortion signal and provide anamplified linear output signal.
 14. The RF transceiver of claim 13,wherein the non-linear circuit is a high power amplifier.
 15. The RFtransceiver of claim 13, wherein the non-linear circuit is a low noiseamplifier.
 16. The RF transceiver of claim 13, further comprising anantenna configured to receive and transmit the linear output signal. 17.The RF transceiver of claim 13, further comprising an antenna configuredto receive the RF signal and provide the RF signal to the adaptivepre-distortion circuit.
 18. The RF transceiver of claim 13, wherein theadaptive pre-distortion circuit comprises: an adder coupled to receivethe feedback signal and a first signal, wherein the adder generates adifference signal between the feedback signal and the first signal,wherein the first signal is a signal to be transmitted that istime-aligned with the feedback signal; an adaptive coefficient generatorconfigured to receive the difference signal and a second signal, whereinthe second signal is a version of the signal to be pre-distorted, andwherein the adaptive coefficient generator generates a set of tapcoefficients; and a timing and filter circuit configured to receive thefeedback signal and the signal to be pre-distorted and generate thefirst signal and receive the tap coefficients and generate thepre-distortion output signal based on a mean square error.
 19. The RFtransceiver of claim 18, further comprising a bandpass filter coupled toreceive the pre-distortion output signal.
 20. The RF transceiver ofclaim 12, wherein the adaptive pre-distortion circuit comprises afeedforward filter having a series of N delay elements, wherein eachdelay element introduces a delay ${\tau = \frac{T_{s}}{N - 1}},$ whereT_(s) is the symbol period.
 21. The RF transceiver of claim 13, whereinthe adaptive pre-distortion circuit comprises an adder that provides adifference signal between the RF signal and the pre-distortion signal,wherein the pre-distortion signal is adaptively changed to minimize thedifference signal.
 22. The RF transceiver of claim 13, furthercomprising a second adaptive pre-distortion circuit configured toreceive an RF signal on a different carrier frequency and the feedbacksignal and adaptively generate a second pre-distortion signal.
 23. TheRF transceiver of claim 22, further comprising a first and a secondbandpass filter coupled to receive the pre-distortion and secondpre-distortion signal, respectively, and an adder configured to receivethe outputs of the first and second bandpass filters.
 24. Apre-distortion circuit, comprising: means for receiving an input signaland a feedback signal from a non-linear circuit; means for determining ametric between the input signal and the feedback signal; means foradaptively changing the input signal based on minimizing the metric; andmeans for generating a non-linear pre-distorted signal using theadaptively changed input signal, wherein the non-linear pre-distortedsignal is input to the non-linear circuit to provide a linear outputsignal.
 25. The pre-distortion circuit of claim 24, wherein the metricis the difference between the input signal and the feedback signal. 26.The pre-distortion circuit of claim 24, wherein the input signal is anRF signal.
 27. A method of pre-distorting an input signal, comprising:receiving an input signal and a feedback signal from a non-linearcircuit; determining a metric between the input signal and the feedbacksignal; adaptively changing the input signal based on minimizing themetric; and generating a non-linear pre-distorted signal using theadaptively changed input signal, wherein the non-linear pre-distortedsignal is input to the non-linear circuit to provide a linear outputsignal.
 28. The method of claim 27, wherein the metric is a differencesignal between the input signal and the feedback signal.
 29. The methodof claim 27, wherein the adaptively changing is in the RF domain.